Method and apparatus for resonant power conversion

ABSTRACT

A method and apparatus for providing multi-phase power. In one embodiment, the apparatus comprises a cycloconverter controller for determining a charge ratio based on a reference waveform; and a cycloconverter, coupled to the cycloconverter controller and to a multi-phase AC line, for selectively coupling an alternating current to each line of the multi-phase AC line based on the charge ratio.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims benefit of U.S. provisional patent applicationSer. No. 61/460,526, filed Jan. 4, 2011, which is herein incorporated inits entirety by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

Embodiments of the present disclosure relate generally to powerconversion, and, in particular, to controlling power conversion in aresonant converter.

2. Description of the Related Art

Resonant converters provide many advantages over other types of powerconverters. Such advantages may include low noise, low component stress,low component count, and predictable conduction-dominated losses.Resonant converters may therefore be smaller, less costly, and moreefficient devices than other types of converters.

Therefore, there is a need in the art for a method and apparatus forefficiently converting a DC voltage to an AC voltage utilizing aresonant converter.

SUMMARY OF THE INVENTION

Embodiments of the present invention generally relate to a method andapparatus for providing multi-phase power. In one embodiment, theapparatus comprises a cycloconverter controller for determining a chargeratio based on a reference waveform; and a cycloconverter, coupled tothe cycloconverter controller and to a multi-phase AC line, forselectively coupling an alternating current to each line of themulti-phase AC line based on the charge ratio.

BRIEF DESCRIPTION OF THE DRAWINGS

So that the manner in which the above recited features of the presentinvention can be understood in detail, a more particular description ofthe invention, briefly summarized above, may be had by reference toembodiments, some of which are illustrated in the appended drawings. Itis to be noted, however, that the appended drawings illustrate onlytypical embodiments of this invention and are therefore not to beconsidered limiting of its scope, for the invention may admit to otherequally effective embodiments.

FIG. 1 is a block diagram of a resonant converter in accordance with oneor more embodiments of the present invention;

FIG. 2 is a block diagram of a bridge controller in accordance with oneor more embodiments of the present invention;

FIG. 3 is a block diagram of a cycloconverter controller in accordancewith one or more embodiments of the present invention;

FIG. 4 is a graph depicting a slice of a three-phase reference currentwaveform in accordance with one or more embodiments of the presentinvention;

FIG. 5 is a set of graphs depicting current generated on each line of athree-phase AC line during a slice in accordance with one or moreembodiments of the present invention;

FIG. 6 is a schematic diagram of an alternative embodiment of acycloconverter;

FIG. 7 is a flow diagram of a method for modulating output power from aresonant power converter in accordance with one or more embodiments ofthe present invention;

FIG. 8 is a flow diagram of a method of operation of an AC currentswitching stage in accordance with one or more embodiments of thepresent invention; and

FIG. 9 is a block diagram of a grid interface controller 156 inaccordance with one or more embodiments of the present invention.

DETAILED DESCRIPTION

FIG. 1 is a block diagram of a resonant converter 100 in accordance withone or more embodiments of the present invention. This diagram onlyportrays one variation of the myriad of possible system configurations.The present invention can function in a variety of power generationenvironments and systems.

The resonant converter 100 comprises a bridge 102 coupled across aparallel input capacitor 130 and a series combination of a capacitor116, an inductor 118, a primary winding 106P of a transformer 106, and acurrent sampler 112. Such components form a DC voltage switching stage160 of the resonant converter 100. In some embodiments, at least aportion of the capacitance of the parallel input capacitor 130 may bedue to parasitic capacitance from switching devices within the resonantconverter 100.

The bridge 102 is a full H-bridge comprising switches 120-1, 120-2,122-1, and 122-2 (e.g., n-type metal-oxide-semiconductor field-effecttransistors, or MOSFETs) arranged such that switches 120-1/120-2 and122-1/122-2 form first and second diagonals, respectively, of theH-bridge. Gate terminals and source terminals of each of the switches120-1, 120-2, 122-1, and 122-2 are coupled to a bridge controller 114for operatively controlling the switches. In other embodiments, theswitches 120-1, 120-2, 122-1, and 122-2 may be any other suitableelectronic switch, such as insulated gate bipolar transistors (IGBTs),bipolar junction transistors (BJTs), p-type MOSFETs, gate turnoffthyristors (GTOs), and the like. The bridge 102 operates at a switchingspeed of approximately 100 kilohertz (kHz) and is able to switch, forexample, from 60 to 600 volts depending upon the DC voltage source tothe bridge; in other embodiments, the bridge 102 may operate at adifferent switching frequency. In some other embodiments, the bridge 102may be a half H-bridge rather than a full H-bridge.

A first output terminal of the bridge 102 is coupled between theswitches 120-1 and 122-2, and is also coupled to a first terminal of theparallel input capacitor 130 and to a first terminal of the capacitor116. A second terminal of the capacitor 116 is coupled to a firstterminal of the inductor 118, and a second terminal of the inductor 118is coupled to a first terminal of the primary winding 106P. Thecapacitor 116 and the inductor 118 form a series resonant circuit 104operating at a frequency of 100 kHz; alternatively, the resonant circuit104 may operate at a different resonant frequency. In some embodiments,the inductor 118 may represent a leakage inductance of the transformer106 rather than being a separate discrete inductor and the resonantcircuit for the converter is formed between the transformer 106 and thecapacitor 116, thereby reducing the overall component count of theresonant converter 100. In other embodiments, other types of resonantcircuits (e.g., series LC, parallel LC, series-parallel LLC,series-parallel LCC, series-parallel LLCC, and the like) may be utilizedwithin the resonant converter 100 in place of or in addition to theresonant circuit 104.

The current sampler 112 is coupled between a second terminal of theprimary winding 106P and a second output terminal of the bridge 102,where the second output terminal is coupled between the switches 122-1and 120-2. Additionally, a voltage sampler 138 is coupled across theparallel input capacitor 130; both the voltage sampler 138 and thecurrent sampler 112 are coupled to a power calculator 140, and the powercalculator 140 is coupled to the bridge controller 114.

On the secondary side of the transformer 106, a first terminal of asecondary winding 106S is coupled to a first terminal of a capacitor108. A second terminal of the capacitor 108 is coupled to a firstterminal of a parallel output capacitor 132, and a second terminal ofthe parallel output capacitor 132 is coupled to a second terminal of thesecondary winding 106S. A cycloconverter 110 is coupled across theparallel output capacitor 132 and forms an AC current switching stage162 of the resonant converter 100. By selection of both the parallelinput capacitor 130 and the parallel output capacitor 132, the resonantconverter 100 can be designed to modulate over a wide range of powerwith a relatively small change in switching frequency of the bridge 102.

In some embodiments, the capacitor 116 may be on the order of 25nanofarad (nF), the inductor 118 may be on the order of 100 microhenries(μH), the parallel input capacitor 130 may be on the order of 1 nF, theparallel output capacitor 132 may be on the order of 5 nF, and thetransformer 106 may have a turns ratio of 1:1.5; such embodiments mayhave a frequency range of 150 kilohertz (kHz)—300 kHz. Generally, theseries capacitance of the resonant circuit 104 may be on the order of 25nF. For example, the capacitor 116 may be on the order of 25 nF and thecapacitor 108 may be made extremely large such that it acts as a DCblocking capacitor and does not affect the resonance of the circuit.Alternatively, for a transformer turns ratio of 1:1.5, the capacitor 116may be on the order of 50 nF and the capacitor 108 may be on the orderof 22.2 nF (i.e., the capacitor 108 appears as a 50 nF capacitor inseries with the capacitor 116 as a result of the transformer turnsratio).

The cycloconverter 110 comprises switches 150-1, 150-2, 152-1, 152-2,154-1, and 154-2. Drain terminals of the switches 150-1, 152-1, and154-1 are coupled to the first terminal of the parallel output capacitor132. Source terminals of each switch pair 150-1/150-2, 152-1/152-2, and154-1/154-2 are coupled together (i.e., the source terminals of switches150-1/150-2 are coupled together, the source terminals of switches152-1/152-2 are coupled together, and the source terminals of switches154-1/154-2 are coupled together). Drain terminals of the switches154-2, 152-2, and 150-2 are coupled to first, second, and third outputterminals, respectively, which in turn are coupled to lines L1, L2, andL3, respectively, of a three-phase AC line. Additionally, the secondterminal of the output parallel capacitor 132 is coupled to a neutralline N of the three-phase AC line. In some embodiments, the AC line maybe a commercial power grid system.

Gate and source terminals of each switch 150-1, 150-2, 152-1, 152-2,154-1, and 154-2 are coupled to a cycloconverter controller 142, whichis further coupled to the current sampler 112. The cycloconvertercontroller 142 operates (i.e., activates/deactivates) each of thecycloconverter switches to couple three-phase AC power to the AC line(i.e., a first phase is coupled to line L1, a second phase is coupled toline L2, and a third phase is coupled to line L3). The switch pair150-1/150-2 form a first four-quadrant switch (i.e., a fullybi-directional switch), the switch pair 152-1/152-2 form a secondfour-quadrant switch, and the switch pair 154-1/154-2 form a thirdfour-quadrant switch. In some embodiments, the switches 150-1, 150-2,152-1, 152-2, 154-1, and 154-2 may be n-type MOSFET switches; in otherembodiments, other suitable switches and/or arrangements of switches maybe utilized for the first, the second, and the third four-quadrantswitches.

A line voltage sampler 144 is coupled to the drain terminals of theswitches 150-2, 152-2, and 154-2 (i.e., lines L1, L2, and L3,respectively), as well as to the second terminal of the output parallelcapacitor 132 (i.e., line N). The line voltage sampler 144 is alsocoupled to a grid interface controller 156. The grid interfacecontroller 156 is further coupled to the cycloconverter controller 142,the bridge controller 114, and a power controller 158.

During operation, the bridge 102 receives an input voltage Vin from a DCvoltage source, such as one or more renewable energy sources (e.g.,photovoltaic (PV) modules, wind farms, hydroelectric systems, or thelike), batteries, or any suitable source of DC power. The bridgecontroller 114 alternately activates/deactivates the H-bridge diagonals(i.e., 180° out of phase) to generate a bridge output voltage Vbr thatis a bipolar square wave; in some embodiments, the frequency at whichthe H-bridge diagonals are switched (i.e., the switching frequency) ison the order of 100 kHz. The bridge output voltage Vbr results in asubstantially sinusoidal current Ir through the resonant circuit 104(operating at a frequency of 100 kHz) and the primary winding 106P,thereby inducing an alternating current in the secondary winding 106S.The transformer 106 may be a step-up transformer for increasing thevoltage from the primary to the secondary (for example, for a DC inputgenerated by a PV module, the transformer 106 would generally be astep-up transformer) or, alternatively, a step-down transformer fordecreasing the voltage.

As a result of the current induced in the secondary winding 106S, asubstantially sinusoidal current waveform Ic at a frequency of 100 kHzflows into the cycloconverter 110. The amplitude of the current waveformIc is controlled by the switching frequency of the bridge 102 and can beincreased or decreased by suitably adjusting the switching frequency ofthe H-bridge; i.e., the current (and power) transferred varies as thesignal frequency moves away from the resonant frequency of the resonantcircuit 104. The power controller 158 determines an output powerrequired from the resonant converter 100 and, via a three-phasereference current waveform generated by the grid interface controller156, drives the bridge controller 114 to adjust the H-bridge switchingfrequency to achieve the required output power. In some embodimentswhere the resonant converter 100 receives input power from a PV module,the power controller 158 may determine the resonant converter requiredoutput power such that the PV module is biased at a maximum power point(MPP). In such embodiments, the power controller 158 may be coupled tothe input of the bridge 102 for determining the voltage and currentprovided by the PV module. In other embodiments, the power controller158 may receive commands from an external source to operate at a givenpower and power factor. For example, the resonant converter 100 mayreceive power from a PV module and the power controller 158 may receivea command (e.g., via the grid interface controller 156 or an alternativemeans) from a utility to run at a lower power than the MPP to helpstabilize the grid.

The current sampler 112 samples the current Ir and generates valuesindicative of the sampled current (“current samples”), while the voltagesampler 138 samples the voltage Vbr and generates values indicative ofthe sampled primary side voltage (“primary voltage samples”). Thecurrent sampler 112 and the voltage sampler 138 may perform suchsampling at a rate of 50 MHz, although other sampling rates may be usedby the current sampler 112 and/or the voltage sampler 138. In someembodiments, the current sampler 112 and the voltage sampler 138 eachcomprise an analog-to-digital converter (ADC) for generating the samplesin a digital format.

The current sampler 112 and the voltage sampler 138 respectively couplethe current samples and primary voltage samples to the power calculator140. Based on the current and voltage samples, the power calculator 140computes the generated power level and couples such computed power levelto the bridge controller 114. The bridge controller 114 then comparesthe computed power level to the required output power level and adjuststhe switching frequency to increase or decrease the generated power asneeded.

The cycloconverter 110 selectively couples the received current waveformIc to each phase of the three-phase AC line at the cycloconverteroutput; in some embodiments, the AC line may be a commercial power gridoperating at 60 Hz. In order to selectively couple the relativelyhigh-frequency current Ic to each phase of the lower frequency AC line,a three-phase reference current waveform (also referred to as “referencecurrent waveform”) is generated by the grid interface controller 156based on the required resonant converter output power from the powercontroller 158 and the three-phase AC line voltage as determined fromline voltage samples generated by the line voltage sampler 144. The linevoltage sampler 144 samples the AC line voltage (i.e., the gridvoltage), for example at a rate of 30 kilosamples per second (kSPS), andcouples one or more values indicative of the sampled line voltages(“line voltage samples”) to the grid interface controller 156. In someembodiments, the line voltage sampler 144 comprises an ADC forgenerating the samples in a digital format. Based on the received linevoltage samples, the grid interface controller 156 generates thereference current waveform synchronous with the grid voltage waveformand couples the reference current waveform to the cycloconvertercontroller 142. The reference current waveform ensures that even if thegrid voltage deviates from a sinewave, each output current generated canbe controlled to match the desired output. In the event of the gridvoltage and/or frequency deviating from required operationalspecifications, a supervisory system (not shown) will deactivate theresonant converter 100.

In some embodiments, i.e., for operating with a power factor of 1, thereference current waveform is generated in phase with the line voltage.In other embodiments where reactive power is being produced by theresonant converter 100, e.g., for providing Volt-Ampere-Reactive (VAR)compensation, the reference current waveform is generated out of phasewith the line voltage. In three phase embodiments the reference currentwaveform will be of a three phase form. In some other embodiments, thereference current waveform may be a single-phase or split-phase ACwaveform for coupling generated current to a single-phase or split-phaseAC line.

Consecutive time windows, or “slices”, of the reference current waveformare individually analyzed by the cycloconverter controller 142 togenerate a “charge ratio” for driving the cycloconverter 110 to couplegenerated current to each output phase. Within each slice, the level ofcurrent for each phase of the reference current waveform can berepresented by a single “DC” current value (i.e., one DC current valueper phase) due to the relatively low line frequency with respect to thecurrent Ic. A first DC current may represent the value of the firstphase of the reference current waveform (e.g., the desired current to beinjected into line L1) during the slice, a second DC current mayrepresent the value of the second phase of the reference currentwaveform (e.g., the desired current to be injected into line L2) duringthe slice, and a third DC current may represent the value of the thirdphase of the reference current waveform (e.g., the desired current to beinjected into line L3) during the slice. Given that the desired currentto be injected on a particular line during a switching period is equalto the charge to be injected divided by the switching period, and theswitching period is constant relative to the line frequency, the ratioof the desired current to be injected on each line is equal to the ratioof the charge to be injected on each line (i.e., the charge ratio). Forexample, if the relative values of the desired currents on linesL1/L2/L3 during a slice are 300/−100/−200, i.e., 3/−1/−2, respectively,L1 should receive the entire positive portion of the charge, L2 shouldreceive ⅓ of the negative portion of the charge, and L3 should receive ⅔of the negative portion of the charge. The charge ratio thus indicatesthe relative levels of current to be coupled to, or “steered” into, eachoutput phase during a particular slice. For each slice, thecycloconverter controller 142 determines a charge ratio from thereference current waveform and operates the cycloconverter 110 toselectively couple the generated current to each phase of the AC line inaccordance with the corresponding charge ratio. In some embodiments,each slice may be a fixed width, i.e., duration; in other embodiments,for example embodiments for multi-phase applications, the width of oneor more slices may be varied (e.g., variation may be determined by theposition in the phase). The cycloconverter 110 operates independent ofthe bridge 102; i.e., the bridge 102 controls the amplitude of theoutput current generated, and the cycloconverter 110 controls the ratioof output current steered into each output phase.

As described in more detail further below, the cycloconverter 110 may beoperated such that one or more of the switches 150-1, 150-2, 152-1,152-2, 154-1, and 154-2 remain activated (i.e., “on”) during an entireslice. Such a “minimal transition” switching technique reduces gatedrive voltage requirements as well as stress on one or more of thecycloconverter switches 150-1, 150-2, 152-1, 152-2, 154-1, and 154-2,thereby improving the overall operating efficiency of the resonantconverter 110. Additionally, the switches within the bridge 102 and thecycloconverter 110 may be operated in a zero-voltage switching (ZVS)mode for further improved efficiency. In some embodiments, the resonantconverter 100 may be operated in a ZVS mode for all of the resonantconverter switching devices over the entire operating range.

In one or more other embodiments, the resonant converter 100 mayinterleave two or more power stages, switch among a plurality of modesof operation, and/or employ a burst technique where energy from the DCinput is stored during one or more line voltage cycles and subsequentlycoupled (i.e., “bursted”) to the AC line during a burst period of one ormore line voltage cycles. In some alternative embodiments, such as theembodiment depicted in FIG. 6 as described in detail below, thecycloconverter 110 generates a single-phase AC output that is coupled toa single-phase AC line.

FIG. 2 is a block diagram of a bridge controller 114 in accordance withone or more embodiments of the present invention. The bridge controller114 comprises support circuits 204 and a memory 206, each coupled to acentral processing unit (CPU) 202. The CPU 202 may comprise one or moreconventionally available microprocessors or microcontrollers;alternatively, the CPU 202 may include one or more application specificintegrated circuits (ASICs). The support circuits 204 are well knowncircuits used to promote functionality of the CPU 202. Such circuitsinclude, but are not limited to, a cache, power supplies, clockcircuits, buses, input/output (I/O) circuits, and the like. The bridgecontroller 114 may be implemented using a general purpose computer that,when executing particular software, becomes a specific purpose computerfor performing various embodiments of the present invention.

The memory 206 may comprise random access memory, read only memory,removable disk memory, flash memory, and various combinations of thesetypes of memory. The memory 206 is sometimes referred to as main memoryand may, in part, be used as cache memory or buffer memory. The memory206 generally stores the operating system (OS) 208, if necessary, of thebridge controller 114 that can be supported by the CPU capabilities.

The memory 206 may store various forms of application software, such asa bridge control module 210 for controlling operation of the bridge 102and performing functions related to the present invention. For example,the bridge controller 114 executes the module 210 to use the requiredoutput power (e.g., as determined from the reference current waveform)and the calculated power from the power calculator 140 (i.e., the powergenerated at the output of the bridge 102) to adjust the bridgeswitching frequency above or below a nominal 100 kHz frequency. Forembodiments where a PV module is coupled at the input of the resonantconverter 100, changing the switching frequency of the bridge 102 altersthe load impedance as seen by the PV module to achieve MPP. Furtherdetail on the functionality provided by the bridge controller 114 isdescribed below with respect to FIG. 7.

The memory 206 may additionally store a database 212 for storing datarelated to the operation of the resonant converter 100 and/or thepresent invention.

In other embodiments, the CPU 202 may be a microcontroller comprisinginternal memory for storing controller firmware that, when executed,provides the controller functionality described below with respect toFIG. 7.

FIG. 3 is a block diagram of a cycloconverter controller 142 inaccordance with one or more embodiments of the present invention. Thecycloconverter controller 142 comprises support circuits 304 and amemory 306, each coupled to a central processing unit (CPU) 302. The CPU302 may comprise one or more conventionally available microprocessors ormicrocontrollers; alternatively, the CPU 302 may include one or moreapplication specific integrated circuits (ASICs). The support circuits304 are well known circuits used to promote functionality of the CPU302. Such circuits include, but are not limited to, a cache, powersupplies, clock circuits, buses, input/output (I/O) circuits, and thelike. The cycloconverter controller 142 may be implemented using ageneral purpose computer that, when executing particular software,becomes a specific purpose computer for performing various embodimentsof the present invention.

The memory 306 may comprise random access memory, read only memory,removable disk memory, flash memory, and various combinations of thesetypes of memory. The memory 306 is sometimes referred to as main memoryand may, in part, be used as cache memory or buffer memory. The memory306 generally stores the operating system (OS) 308, if necessary, of thecycloconverter controller 142 that can be supported by the CPUcapabilities.

The memory 306 may store various forms of application software, such asa cycloconverter control module 310 for controlling operation of thecycloconverter 110 and performing functions related to the presentinvention. For example, the cycloconverter control module 310 monitorsthe high frequency current, determines the charge ratio for each slice,compares parameters (e.g., values of the line voltage phases) todetermine whether any “dead zones” exist as described below, andselectively couples each generated current pulse to appropriate lines ofthe AC line based on the charge ratio; in some alternative embodiments,the grid interface controller 156 may compare relevant parameters todetermine whether any dead zones exist. In some embodiments, thecycloconverter control module 310 may compute one or more slice widthsbased on one or more stored algorithms. Further detail on thefunctionality provided by the cycloconverter control module 310 isdescribed below with respect to FIG. 8.

The memory 306 may additionally store a database 312 for storing datarelated to the operation of the cycloconverter 110 and/or the presentinvention, such as one or more dead zone thresholds, one or more DCvoltages, one or more predetermined slice widths, one or more algorithmsfor determining slice widths, or the like.

In other embodiments, the CPU 302 may be a microcontroller comprisinginternal memory for storing controller firmware that, when executed,provides the controller functionality described below with respect toFIG. 8.

In some embodiments, the bridge controller 114 and the cycloconvertercontroller 142 may be a single controller controlled by the same CPU;i.e., a single controller may execute both the bridge control module 210and the cycloconverter control module 310.

FIG. 4 is a graph 400 depicting a slice 410 of a three-phase referencecurrent waveform 408 in accordance with one or more embodiments of thepresent invention. The graph 400 depicts a first phase waveform 402 ofthe reference current waveform 408 (e.g., a desired current to beinjected into line L1), a second phase waveform 404 of the referencecurrent waveform 408 (e.g., a desired current to be injected into lineL2), and a third phase waveform 406 of the reference current waveform408 (e.g., a desired current to be injected into line L3). The waveforms402, 404, and 406 form the three-phase reference current waveform 408;for example, the waveforms 402, 404, and 406 each are at a frequency of60 Hz and are offset from one another by 120 degrees. In someembodiments, the three-phase reference current waveform 408 is areference for a desired current to be coupled to a commercial powergrid.

A time window across the three-phase reference current waveform 408(i.e., across each waveform 402, 404, and 406) is shown as the slice410. The slice 410 starts at a start time TS and ends at an end time TE.In some embodiments, the width of the slice 410 (i.e., the time from TSto TE) may be approximately three-orders of magnitude faster than thethree-phase reference current waveform 408. For example, for acommercial power grid coupled to the resonant converter 100 andoperating at a frequency of 60 Hz, the slice 410 may have a width (i.e.,duration) on the order of 10 s of microseconds. In other embodiments,the width of the slice 410 may be greater or less than 10 microseconds.

At time TS, the waveform 402 has a value of DC1, the waveform 404 has avalue of DC2, and the waveform 406 has a value of DC3. The values DC1,DC2, and DC3 may be used as DC current values to represent the values ofthe waveforms 402, 404, and 406, respectively, during the entire slice410 (i.e., from time TS to time TE). The ratio of the DC current valuesprovides the charge ratio for the slice 410 for operating thecycloconverter switches from the time TS to the time TE, as described indetail below with respect to FIG. 5.

FIG. 5 is a set of graphs 500 depicting current selectively coupled intoeach line of a three-phase AC line during a slice 410 in accordance withone or more embodiments of the present invention. As previouslydescribed, the cycloconverter controller 142 operates the cycloconverter110 during a slice to selectively couple, or steer, current into eachphase of the AC line in accordance with the charge ratio for the slice.For each cycle of the high-frequency current Ic during the slice 410,the cycloconverter 110 divides the current Ic based on the charge ratio(i.e., DC1/DC2/DC3) and selectively couples the divided current to theappropriate output line. Accordingly, the cycloconverter controller 142does not require information pertaining to the actual values of currentto be steered into each line, but only the relative ratios of thecurrents.

The graphs 500 comprise a first graph 502 depicting current steered intoline L1 during the slice 410, a second graph 504 depicting currentsteered into line L2 during the slice 410, and a third graph 506depicting current steered into line L3 during the slice 410. In someembodiments, such as the embodiment depicted in FIG. 5, the relativevalues of DC1, DC2, and DC3 for the slice 410 may be 3/−1/−2,respectively. In accordance with the charge ratio, the cycloconverter110 selectively couples the entire positive portion of the charge toline L1, ⅓ of the negative portion of the charge to line L2, and ⅔ ofthe negative portion of the charge to line L3 as described below.

From time TS to T1, the cycloconverter controller 142 activates theswitch pair 154-1/154-2 to couple the positive portion of the current Icto line L1. Due to parasitic diodes of the switches, only one switch ineach switch pair 150-1/150-2 and 152-1/152-2 needs to be deactivatedunder certain operating conditions to prevent any current from beingcoupled to lines L2 or L3; such a minimal transition switching techniqueof deactivating one switch within a pair while leaving the remainingswitch active during the current cycle reduces energy requirements forswitching in the resonant converter 100. To use minimal transitionswitching, the line voltage values on those lines not receiving outputcurrent must be sufficiently separated in order to prevent direct shortsbetween phases that may occur if all corresponding switches are notdeactivated, If the difference between the voltages is equal to orexceeds a “dead zone” threshold, minimal transition switching may beused for the corresponding switch pairs during the current cycle; if thedifference between the voltages is less than the dead zone threshold,minimal transition switching is not used for the corresponding switchpairs during the current cycle.

For the line voltages corresponding to the second and third phasewaveforms 404 and 406 during the slice 410 (e.g., the values of the linevoltages on lines L2 and L3 at time TS), if the values are close enoughto satisfy a dead zone threshold, all of the switches in the switchpairs 150-1/150-2 and 152-1/152-2 are deactivated to prevent a directshort between the two phase lines L2 and L3. If, however, the values ofthe line voltages corresponding to the second and third phase waveforms404 and 406 during the slice 410 are not close enough to satisfy thedead zone threshold, as in the embodiment depicted in FIG. 4, minimaltransition switching may be employed and only one switch in each switchpair 150-1/150-2 and 152-1/152-2 need be deactivated. In someembodiments, the dead zone threshold may be on the order of 20 volts.

As a result of such switch activation/deactivation, the current steeredinto line L1 from time TS to T1 is a first half-cycle of the currentwaveform Ic; i.e., 100% of the positive portion of the Ic cycle asdictated by the charge ratio.

From time T1 to T2, the cycloconverter controller 142 activates theswitch pair 152-1/152-2 and deactivates at least one switch in eachswitch pair 150-1/150-2 and 154-1/154-2 based on whether the values ofthe line voltages corresponding to the first and third phase waveforms402 and 406 (e.g., the values of the line voltages on lines L1 and L3 attime TS) are close enough during the slice 410 to satisfy the dead zonethreshold. For the embodiment depicted in FIG. 4, the dead zonethreshold is not satisfied and only one switch in each switch pair150-1/150-2 and 154-1/154-2 is deactivated (i.e., minimal transitionswitching is used). As a result of such switch activation/deactivation,the current steered into line L2 from time T1 to T2 is ⅓ of a secondhalf-cycle of the current waveform Ic; i.e., 33.3% of the negativeportion of the Ic cycle as dictated by the charge ratio.

From time T2 to T3, the cycloconverter controller 142 activates theswitch pair 150-1/150-2 and deactivates at least one switch in eachswitch pair 152-1/152-2 and 154-1/154-2 based on whether the values ofthe line voltages corresponding to the first and second phase waveforms402 and 404 (e.g., the values of the line voltages on lines L1 and L2 attime TS) are close enough during the slice 410 to satisfy the dead zonethreshold. For the embodiment depicted in FIG. 4, the dead zonethreshold is not satisfied and only one switch in each switch pair152-1/152-2 and 154-1/154-2 is deactivated (i.e., minimal transitionswitching is used). As a result of such switch activation/deactivation,the current steered into line L3 from time T2 to T3 is ⅔ of the secondhalf-cycle of the current waveform Ic; i.e., 66.7% of the negativeportion of the Ic cycle as dictated by the charge ratio. Thus, from timeTS to time T3, one full cycle of the current Ic is selectively coupledto the lines L1, L2, and L3.

From time T3 to TE, one or more additional cycles of the currentwaveform Ic may be selectively coupled to the lines L1, L2, and L3 inthe same manner as during the time TS to T3 (i.e., in accordance withthe charge ratio for the slice 410). In some embodiments, the width ofthe slice 410 is 10 microseconds and a single cycle of the currentwaveform Ic is transferred to the lines L1, L2, and L3 during the slice410.

By selectively coupling the current Ic to the lines L1, L2, and L3 asdescribed above, i.e., by partitioning each individual cycle of thecurrent Ic based on the charge ratio and coupling the current Ic tooutput lines L1, L2, and L3 accordingly and by using minimal transitionswitching, the switch pairs 150-1/150-2 and 154-1/154-2 havezero-voltage switching (ZVS) and zero-current switching (ZCS)transitions, while the switch pair 152-1/152-2 has a ZVS transition atturn-on but no ZCS transition. As such, only one switch pair within thecycloconverter 110 experiences any switching losses and thecycloconverter 110 has the most zero-loss transitions possible.

Additionally, partitioning each individual cycle of the current Ic basedon the charge ratio leads to the lowest peak-to-peak current waveformthat can possibly be obtained on the cycloconverter 110 to obtain thedesired power output. Further, such operation ensures that the voltagewaveform being placed at the input is the most in-phase with the currentthat can possibly be obtained, making the load look as resistive aspossible and improving system stability.

In one or more other embodiments, the times T1, T2, and/or T3 may belonger or shorter; i.e., each switch pair 150-1/150-2, 152-1/152-2, and154-1/154-2 may remain on for a longer or shorter duration toselectively couple current to the lines L1, L2, and L3, although minimalswitching losses are no longer experienced (i.e., ZVS/ZCS cannot beemployed). In such embodiments, the ratio of current steered into eachleg remains defined by the charge ratio for the slice but eachindividual cycle of the current Ic is not partitioned as per the chargeratio. For example, in an alternative embodiment having the charge ratio3/−1/−2, the cycloconverter switches may be operated such that a firstIc cycle within a slice has all of its negative current steered intoline L2 and subsequent second and third Ic cycles within the slice haveall of their negative current steered into the line L3.

FIG. 6 is a schematic diagram of an alternative embodiment of acycloconverter 110. The cycloconverter 110 depicted in FIG. 6 may beused for a single-phase application and comprises switches 602-1, 602-2,604-1, and 604-2. A drain terminal of the switch 602-1 is coupled to thefirst terminal of the parallel output capacitor 132 and to a sourceterminal of the switch 604-1. Source terminals of the switches 602-1 and602-2 are coupled together, and a drain terminal of the switch 602-2 iscoupled to the second terminal of the output parallel capacitor 132 anda source terminal of the switch 604-2. Gate terminals and sourceterminals of each switch 602-1, 602-2, 604-1, and 604-2 are coupled tothe cycloconverter controller 142 for operating (i.e.,activating/deactivating) each of the switches. The switch pair602-1/602-2 forms a first four-quadrant switch and the switch pair604-1/604-2 forms a second four-quadrant switch. In some embodiments,the switches 602-1, 602-2, 604-1, and 604-2 may be n-type MOSFETswitches; in other embodiments, other suitable switches and arrangementsof switches may be utilized for the first and the second four-quadrantswitches.

Drain terminals of the switches 604-1 and 604-2 are coupled to first andsecond output terminals, respectively, which in turn are coupled tolines L1 and N of a single-phase AC line. The line voltage sampler 144is coupled across the first and second output terminals for sampling theAC line voltage.

During operation, the cycloconverter controller 142 operates theswitches to half-wave rectify the current Ic into line L1 and steer theremaining half-period into the line N based on a single-phase currentreference waveform received from the grid interface controller 156. Whenthe single-phase current reference waveform is positive, thecycloconverter controller 142 activates the switch pair 604-1/604-2 anddeactivates the switch pair 602-1/602-2 (i.e., one or both of switches602-1/602-2 are deactivated) during each positive half-cycle of thecurrent Ic; during each negative half-cycle of the current Ic, thecycloconverter controller 142 deactivates the switch pair 604-1/604-2(i.e., one or both of switches 604-1/604-2 are deactivated) andactivates the switch pair 602-1/602-2. Through such operation of thecycloconverter switches, the entire positive portion of the charge isinjected into line L1 and the entire negative portion of the charge isinjected into neutral.

When the single-phase current reference waveform is negative, thecycloconverter controller 142 activates the switch pair 604-1/604-2 anddeactivates the switch pair 602-1/602-2 (i.e., one or both of switches602-1/602-2 are deactivated) during each negative half-cycle of thecurrent Ic; during each positive half-cycle of the current Ic, thecycloconverter controller 142 deactivates the switch pair 604-1/604-2(i.e., one or both of switches 604-1/604-2 are deactivated) andactivates the switch pair 602-1/602-2 during each positive half-cycle ofthe current Ic. Through such operation of the cycloconverter switches,the entire negative portion of the charge is injected into line L1 andthe entire positive portion of the charge is injected into neutral.

During such a single phase application, both switching transitions arelossless.

FIG. 7 is a flow diagram of a method 700 for modulating output powerfrom a resonant power converter in accordance with one or moreembodiments of the present invention. The method 700 is animplementation of the bridge controller 114.

In some embodiments, such as the embodiment described below, theresonant converter (e.g., the resonant converter 100) is coupled to aphotovoltaic (PV) module for receiving a DC input voltage. The resonantconverter utilizes a full-bridge within a DC-DC voltage switching stageat the input of the converter to generate a square wave from the DCinput voltage. The resonant converter then converts the square wave toan AC output voltage. In one or more alternative embodiments, theresonant converter may utilize a half-bridge rather than a full-bridgefor generating a square wave at the input of the converter.

The method 700 begins at step 702 and proceeds to step 704. At step 704,a required output power from the resonant converter is determined forbiasing the PV module at a maximum power point (MPP). In someembodiments, a power controller, such as power controller 158, maydetermine the appropriate resonant converter output power. In one ormore alternative embodiments, the resonant converter may be coupled to aDC power source other than a PV module, and a different required outputpower may be determined. At step 706, a switching frequency of thebridge is determined that will result in the required output power;i.e., the frequency is determined to produce the proper load impedanceto the PV module to obtain maximum power from the PV module at currentoperating conditions. A bridge controller (such as the bridge controller114) may determine the switching frequency based on the output powerrequirements and operate the bridge in accordance with the switchingfrequency. In one or more embodiments, output power requirements may beprovided, e.g., to the bridge controller, via a reference currentwaveform that indicates a desired current to be generated by theresonant converter, where the reference current waveform is generatedfrom one or more samples of an AC line voltage. In some embodiments, theswitching frequency of the bridge may be on the order of 100 kilohertz(kHz) (i.e., the resonant frequency) to achieve the desired outputpower.

The method 700 proceeds to step 708 where the bridge is operated at thedetermined switching frequency. At step 710, output power from thebridge is monitored. For example, a current sampler (e.g., the currentsampler 112) and a voltage sampler (e.g., the voltage sampler 138) mayobtain current and voltage samples, respectively, of the current andvoltage levels generated by the bridge. Such current and voltagessamples are then utilized to compute the power from the bridge.

At step 712, a decision is made whether the power from the bridge shouldbe modified (increased or decreased) in order to meet the converteroutput power requirement for MPP. In some embodiments, the bridgecontroller may receive the computed bridge power and make such adecision. If, at step 712, it is decided that the power from the bridgemust be adjusted, the method 700 returns to step 706 where a newswitching frequency is determined based on whether the bridge power mustbe increased or decreased. Such a feedback loop is performed tocontinuously optimize the output power of the DC-DC switching stage ofthe resonant converter. If, at step 712, it is decided that the bridgepower does not require any modification, the method 700 proceeds to step714.

At step 714, a decision is made whether to continue operating theresonant converter. If, at step 714, it is decided to continueoperation, the method 700 returns to step 708. If, at step 714, it isdecided that operation will not continue, the method 700 proceeds tostep 716 where it ends.

FIG. 8 is a flow diagram of a method 800 of operation of an AC currentswitching stage of a resonant converter in accordance with one or moreembodiments of the present invention. In some embodiments, the method800 is an implementation of the cycloconverter controller 142. In one ormore embodiments, such as the embodiment described below, the AC currentswitching stage comprises a cycloconverter comprising threefour-quadrant switches and is part of a resonant converter that convertsa DC input to a three-phase AC output (e.g., the AC current switchingstage 162 and cycloconverter 110 depicted in FIG. 1). In otherembodiments, the AC current switching stage comprises a cycloconvertercomprising two four-quadrant switches and is part of a resonantconverter that converts the DC input to a single-phase AC output (e.g.,the AC current switching stage 162 and the cycloconverter 110 depictedin FIG. 6).

The method 800 starts at step 802 and proceeds to step 804. At step 804,a slice of a three-phase reference current waveform is determined. Thereference current waveform is a three-phase AC waveform and indicates adesired current to be coupled to an AC line at the output of theresonant converter, such as a commercial AC power grid operating at 60Hz. The reference current waveform is synchronous with a line voltagewaveform on the AC line and is generated based on samples of the linevoltage. In some embodiments, i.e., for operating with a power factor of1, the reference current waveform is generated in phase with the linevoltage waveform. In other embodiments where reactive power is beingproduced by the resonant converter, e.g., for providing VARcompensation, the reference current waveform is generated out of phasewith the line voltage waveform as required.

As previously described, the slice of the reference current waveform isa time window across all three phases of the reference current waveform.The slice may have a fixed width, for example on the order of 10microseconds when the AC line is at 60 Hz, or the width may be variablefrom slice to slice (e.g., variation may be determined by the positionin the phase). In some embodiments, a grid interface controller (e.g.,grid interface controller 156) generates the reference current waveformbased on inputs from a line voltage sampler and a power controller(e.g., line voltage sampler 144 and power controller 158), and couplesthe reference current waveform to a cycloconverter controller (e.g.,cycloconverter controller 142) for determining the slice. At step 806,first, second, and third DC current values are determined to representthe current levels of the first, second, and third phases, respectively,of the reference current waveform within the slice. In some embodiments,the values of the current reference waveform phases at the beginning ofthe slice are utilized as the DC current values.

The method 800 proceeds to step 808, where a charge ratio for the sliceis determined based on the DC values (i.e., the ratio of the first, thesecond, and the third DC current values). At step 810, the levels of theline voltage phases corresponding to the reference current waveformphases within the slice are compared to determine whether any are closeenough in value for the slice to be considered a dead zone. If any twoof the line voltage phases are close enough during the slice to satisfya dead zone threshold (e.g., a threshold of 20V), it is determined thata dead zone exists for the corresponding lines. If no two of the linevoltage phases are close enough during the slice to satisfy thethreshold, it is determined that no dead zone exists. In someembodiments, the values of the line voltages at the beginning of theslice may be compared for determining any dead zones.

At step 812, a decision is made whether a dead zone exists. If no deadzone exists, the method 800 proceeds to step 814 and the cycloconverteris operated during the slice using minimal transition switching, aspreviously described. If a dead zone does exist, the method 800 proceedsto step 816 and the cycloconverter is operated without using minimaltransition switching for the corresponding switches, as previouslydescribed. At both of steps 814 and 816, an approximately sinusoidalcurrent, such as the current Ic, is generated at the input to thecycloconverter. The input current has a high frequency as compared tothe AC line frequency; e.g., the cycloconverter may be coupled to an ACpower grid operating at 60 Hz, and the input current may be on the orderof 100 KHz. The cycloconverter divides the high-frequency input currentbased on the charge ratio for the slice and selectively couples thedivided current to the appropriate output lines.

The method 800 proceeds from each of steps 814 and 816 to step 818. Atstep 818, a decision is made whether or not to continue operating theresonant converter. If a decision is made to continue, the method 800returns to step 804 to determine the next slice; alternatively, if adecision is made to not continue, the method 800 proceeds to step 820where it ends.

FIG. 9 is a block diagram of a grid interface controller 156 inaccordance with one or more embodiments of the present invention. Thegrid interface controller 156 comprises support circuits 904 and amemory 906, each coupled to a central processing unit (CPU) 902. The CPU902 may comprise one or more conventionally available microprocessors ormicrocontrollers; alternatively, the CPU 902 may include one or moreapplication specific integrated circuits (ASICs). The support circuits904 are well known circuits used to promote functionality of the CPU902. Such circuits include, but are not limited to, a cache, powersupplies, clock circuits, buses, input/output (I/O) circuits, and thelike. The grid interface controller 156 may be implemented using ageneral purpose computer that, when executing particular software,becomes a specific purpose computer for performing various embodimentsof the present invention.

The memory 906 may comprise random access memory, read only memory,removable disk memory, flash memory, and various combinations of thesetypes of memory. The memory 906 is sometimes referred to as main memoryand may, in part, be used as cache memory or buffer memory. The memory906 generally stores the operating system (OS) 908, if necessary, of thegrid interface controller 156 that can be supported by the CPUcapabilities.

The memory 906 may store various forms of application software, such asa grid interface control module 910 for performing functions related tothe present invention. For example, the grid interface control module910 may generate the single-phase reference current waveform and/or thethree-phase reference current waveform, synchronize the single-phasereference current waveform and/or the three-phase reference current withan AC line voltage (for example, via a phase-locked loop), and the like.

The memory 906 may additionally store a database 912 for storing datarelated to the present invention, such as required output power levels,or the like.

In other embodiments, the CPU 902 may be a microcontroller comprisinginternal memory for storing controller firmware that, when executed,provides the grid interface controller functionality.

In some embodiments, two or more of the bridge controller 114, thecycloconverter controller 142, and the grid interface controller 156 maybe a single controller controlled by the same CPU; i.e., a singlecontroller may execute two or more of the bridge control module 210, thecycloconverter control module 310, or the grid interface control module910.

The foregoing description of embodiments of the invention comprises anumber of elements, devices, circuits and/or assemblies that performvarious functions as described. For example, the cycloconverter is anexample of a means for selectively coupling an alternating current toeach line of the multi-phase AC line and the cycloconverter controlleris an example of a means for determining a charge ratio from a referencewaveform and driving the cycloconverter to selectively couple thealternating current to each line of the multi-phase AC line based on acharge ratio. These elements, devices, circuits, and/or assemblies areexemplary implementations of means for performing their respectivelydescribed functions.

While the foregoing is directed to embodiments of the present invention,other and further embodiments of the invention may be devised withoutdeparting from the basic scope thereof, and the scope thereof isdetermined by the claims that follow.

The invention claimed is:
 1. An apparatus for providing multi-phasepower, comprising: a transformer comprising a primary winding and asecondary winding; a resonant circuit coupled to the primary winding; acycloconverter controller for determining a charge ratio based on areference waveform; and a cycloconverter, coupled to the secondarywinding, to the cycloconverter controller and to a multi-phase AC line,for selectively coupling an alternating current to each line of themulti-phase AC line based on the charge ratio, wherein the alternatingcurrent results from the resonant circuit and is a sinusoidally shapedsingle-phase waveform; and wherein selectively coupling the alternatingcurrent comprises partitioning a cycle of the alternating current basedon the charge ratio and coupling each partitioned portion of thealternating current to a different phase of the multi-phase AC line. 2.The apparatus of claim 1, wherein the charge ratio indicates relativevalues of each phase of the reference waveform during a time window. 3.The apparatus of claim 2, wherein the cycloconverter controllerdynamically determines the charge ratio for each time window of aplurality of consecutive time windows.
 4. The apparatus of claim 1,wherein the reference waveform is synchronous with an AC voltage on themulti-phase AC line.
 5. The apparatus of claim 1, wherein the resonantcircuit is a series resonant circuit comprising a leakage inductance ofthe transformer and a capacitor.
 6. The apparatus of claim 1, whereinselectively coupling the alternating current comprises deactivating asingle switch on each output line of at least one output line.
 7. Amethod for providing multi-phase power, comprising: generating analternating current via a resonant circuit coupled to a primary windingof a transformer; coupling the alternating current to a cycloconverercoupled to a secondary winding of the transformer; determining a chargeratio based on a reference waveform; selectively coupling, by thecycloconverter, the alternating current to each line of a multi-phase ACline based on the charge ratio, wherein the alternating current is asinusoidally shaped single-phase waveform; wherein selectively couplingthe alternating current comprises partitioning a cycle of thealternating current based on the charge ratio and coupling eachpartitioned portion of the alternating current to a different phase ofthe multi-phase AC line.
 8. The method of claim 7, wherein the chargeratio indicates relative values of each phase of the reference waveformduring a time window.
 9. The method of claim 8, wherein the charge ratiois dynamically determined for each time window of a plurality ofconsecutive time windows.
 10. The method of claim 7, further comprisinggenerating the reference waveform synchronous with an AC voltage on themulti-phase AC line.
 11. The method of claim 7, wherein the resonantcircuit is a series resonant circuit comprising a leakage inductance ofthe transformer and a capacitor.
 12. The method of claim 7, whereinselectively coupling the alternating current comprises deactivating asingle switch on each output line of at least one output line.
 13. Asystem for providing multi-phase power, comprising: a photovoltaic (PV)module; and a resonant converter, coupled to the PV module and to amulti-phase AC line, comprising: a transformer comprising a primarywinding and a secondary winding; a resonant circuit coupled to theprimary winding; a cycloconverter controller for determining a chargeratio based on a reference waveform; a cycloconverter, coupled to thecycloconverter controller and to the multi-phase AC line, forselectively coupling an alternating current to each line of themulti-phase AC line based on the charge ratio, wherein the alternatingcurrent results from the resonant circuit and is a sinusoidally shapedsingle-phase waveform; wherein selectively coupling the alternatingcurrent comprises partitioning a cycle of the alternating current basedon the charge ratio and coupling each partitioned portion of thealternating current to a different phase of the multi-phase AC line. 14.The system of claim 13, wherein the charge ratio indicates relativevalues of each phase of the reference waveform during a time window. 15.The system of claim 13, wherein the reference waveform is synchronouswith an AC voltage on the multi-phase AC line.
 16. The system of claim13, wherein the resonant circuit is a series resonant circuit comprisinga leakage inductance of the transformer and a capacitor.
 17. The systemof claim 13, wherein selectively coupling the alternating currentcomprises deactivating a single switch on at least one output line.